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agilent atf-521p8 high linearity enhancement mode [1] pseudomorphic hemt in 2x2 mm 2 lpcc [3] package data sheet description agilent technologiess atf-521p8 is a single-voltage high linearity, low noise e-phemt housed in an 8-lead jedec- standard leadless plastic chip carrier (lpcc [3] ) package. the device is ideal as a medium-power, high-linearity am- plifier. its operating frequency range is from 50 mhz to 6 ghz. the thermally efficient package mea- sures only 2mm x 2mm x 0.75mm. its backside metalization provides excel- lent thermal dissipation as well as vi- sual evidence of solder reflow. the device has a point mttf of over 300 years at a mounting temperature of +85 c. all devices are 100% rf & dc tested. features ? single voltage operation ? high linearity and p1db ? low noise figure ? excellent uniformity in product specifications ? small package size: 2.0 x 2.0 x 0.75 mm 3 ? point mttf > 300 years [2] ? msl-1 and lead-free ? tape-and-reel packaging option available specifications 2 ghz; 4.5v, 200 ma (typ.) ? 42 dbm output ip3 ? 26.5 dbm output power at 1 db gain compression ? 1.5 db noise figure ? 17 db gain ? 12.5 db lfom [4] applications ? front-end lna q2 and q3, driver or pre-driver amplifier for cellular/ pcs and wcdma wireless infrastructure ? driver amplifier for wlan, wll/rll and mmds applications ? general purpose discrete e-phemt for other high linearity applications pin connections and package marking note: package marking provides orientation and identification 2p = device code x = month code indicates the month of manufacture. note: 1. enhancement mode technology employs a single positive v gs , eliminating the need of negative gate voltage associated with conventional depletion mode devices. 2. refer to reliability datasheet for detailed mttf data 3. conform to jedec reference outline mo229 for drp-n 4. linearity figure of merit (lfom) is essentially oip3 divided by dc bias power. pin 1 (source) pin 2 (gate) pin 3 pin 4 (source) pin 8 pin 7 (drain) pin 6 pin 5 2px top view pin 8 source (thermal/rf gnd) pin 7 (drain) pin 6 pin 5 pin 1 (source) pin 2 (gate) pin 3 pin 4 (source) bottom view
2 atf-521p8 absolute maximum ratings [1] absolute symbol parameter units maximum v ds drain C source voltage [2] v7 v gs gate C source voltage [2] v -5 to 1 v gd gate drain voltage [2] v -5 to 1 i ds drain current [2] ma 500 i gs gate current ma 46 p diss total power dissipation [3] w 1.5 p in max. rf input power dbm 27 t ch channel temperature c 150 t stg storage temperature c -65 to 150 ch_b thermal resistance [4] c/w 45 notes: 1. operation of this device in excess of any one of these parameters may cause permanent damage. 2. assumes dc quiescent conditions. 3. board (package belly) temperaturet b is 25 c. derate 22 mw/ c for t b > 83 c. 4. channel to board thermal resistance measured using 150 c liquid crystal measurement method. product consistency distribution charts [5, 6] nf (db) figure 2. nf @ 2 ghz, 4.5 v, 200 ma. nominal = 1.5 db. 0 1.5 0.5 1 2.5 23 180 150 120 90 60 30 0 stdev = 0.19 +3 std -3 std oip3 (dbm) figure 3. oip3 @ 2 ghz, 4.5 v, 200 ma. nominal = 41.9 dbm, lsl = 38.5 dbm. 37 43 39 41 47 45 49 150 120 90 60 30 0 cpk = 0.86 stdev = 1.32 +3 std -3 std gain (db) figure 4. gain @ 2 ghz, 4.5 v, 200 ma. nominal = 17.2 db, lsl = 15.5 db, usl = 18.5 db. 15 18 16 17 19 180 150 120 90 60 30 0 cpk = 2.13 stdev = 0.21 +3 std -3 std notes: 5. distribution data sample size is 500 samples taken from 5 different wafers. future wafers allocated to this product may have nominal values anywhere between the upper and lower limits. 6. measurements are made on production test board, which represents a trade-off between optimal oip3, p1db and vswr. circuit losses have been de-embedded from actual measurements. v ds (v) figure 1. typical i-v curves. (v gs = 0.1 v per step) i ds (ma) 0.5v vgs = 0.6v 0.7v 0.8v 0.4v 02 6 48 600 500 400 300 200 100 0 p1db (dbm) figure 5. p1db @ 2 ghz, 4.5 v, 200 ma. nominal = 26.5 dbm, lsl = 25 dbm. 25 26.5 25.5 26 27 27.5 300 250 200 150 100 50 0 cpk = 4.6 stdev = 0.11 +3 std -3 std 5. device can safely handle +27dbm rf input power provided igs is limited to 46ma. igs at p 1db drive level is bias circuit dependent.
3 atf-521p8 electrical specifications t a = 25 c, dc bias for rf parameters is vds = 4.5v and ids = 200 ma unless otherwise specified. symbol parameter and test condition units min. typ. max. vgs operational gate voltage vds = 4.5v, ids = 200 ma v 0.62 vth threshold voltage vds = 4.5v, ids = 16 ma v 0.28 idss saturated drain current vds = 4.5v, vgs = 0v a 14.8 gm transconductance vds = 4.5v, gm = ? idss/ ? vgs; mmho 1300 vgs = vgs1 - vgs2 vgs1 = 0.55v, vgs2 = 0.5v igss gate leakage current vds = 0v, vgs = -4v a -20 0.49 nf noise figure [1] f = 2 ghz db 1.5 f = 900 mhz db 1.2 g gain [1] f = 2 ghz db 15.5 17 18.5 f = 900 mhz db 17.2 oip3 output 3 rd order f = 2 ghz dbm 38.5 42 intercept point [1] f = 900 mhz dbm 42.5 p1db output 1db f = 2 ghz dbm 25 26.5 compressed [1] f = 900 mhz dbm 26.5 pae power added efficiency f = 2 ghz % 45 60 f = 900 mhz % 56 aclr adjacent channel leakage offset bw = 5 mhz dbc -51.4 power ratio [1,2] offset bw = 10 mhz dbc -61.5 notes: 1. measurements obtained using production test board described in figure 6. 2. aclr test spec is based on 3gpp ts 25.141 v5.3.1 (2002-06) C test model 1 C active channels: pccpch + sch + cpich + pich + sccpch + 64 dpch (sf=128) C freq = 2140 mhz C pin = -5 dbm C chan integ bw = 3.84 mhz input 50 ohm transmission line including gate bias t (0.3 db loss) input matching circuit _mag = 0.55 _ang = -166 (1.1 db loss) output matching circuit _mag = 0.35 _ang = 168 (0.9 db loss) dut 50 ohm transmission line and drain bias t (0.3 db loss) output figure 6. block diagram of the 2 ghz production test board used for nf, gain, oip3 , p1db and pae and aclr measurements. this circuit achieves a trade-off between optimal oip3, p1db and vswr. circuit losses have been de-embedded from actual measurements.
4 gamma load and source at optimum oip3 and p1db tuning conditions the devices optimum oip3 and p1db measurements were determined using a maury load pull system at 4.5v, 200 ma quiesent bias: optimum oip3 freq gamma source gamma load oip3 gain p1db pae (ghz) mag ang (deg) mag ang (deg) (dbm) (db) (dbm) (%) 0.9 0.413 10.5 0.314 179.0 42.7 16.0 27.0 54.0 2 0.368 162.0 0.538 -176.0 42.5 15.8 27.5 55.3 2.4 0.318 169.0 0.566 -169.0 42.0 14.1 27.4 53.5 3.9 0.463 -134.0 0.495 -159.0 40.3 9.6 27.3 43.9 optimum p1db freq gamma source gamma load oip3 gain p1db pae (ghz) mag ang (deg) mag ang (deg) (dbm) (db) (dbm) (%) 0.9 0.587 12.7 0.613 -172.1 39.1 14.5 29.3 49.6 2 0.614 126.1 0.652 -172.5 39.5 12.9 29.3 49.5 2.4 0.649 145.0 0.682 -171.5 40.0 12.0 29.4 46.8 3.9 0.552 -162.8 0.670 -151.2 38.1 9.6 27.9 39.1 figure 7. simplified schematic of production test board. primary purpose is to show 15 ohm series resistor placement in gate supply. transmission line tapers, tee intersections, bias lines and parasitic values are not shown. rf input 1.5 pf 3.9 nh 1.5 pf rf outpu t 50 ohm .02 110 ohm .03 110 ohm .03 50 ohm .02 dut 1 pf 12 nh 15 ohm 2.2 f gate supply 47 nh 2.2 f drain supply
5 atf-521p8 typical performance curves (at 25 c unless specified otherwise) tuned for optimal oip3 note: bias current for the above charts are quiescent conditions. actual level may increase depending on amount of rf drive. figure 8. oip3 vs. i ds and v ds at 2 ghz. 4.5v 4v 3v i d (ma) oip3 (dbm) 50 45 40 35 30 25 20 15 10 100 400 200 150 300 350 250 figure 9. oip3 vs. i ds and v ds at 900 mhz. i d (ma) oip3 (dbm) 45 40 35 30 25 20 15 10 100 400 200 150 300 350 250 4.5v 4v 3v figure 10. oip3 vs. i ds and v ds at 3.9 ghz. i d (ma) oip3 (dbm) 50 45 40 35 30 25 20 15 10 100 400 200 150 300 350 250 4.5v 4v 3v figure 11. p1db vs. i dq and v ds at 2 ghz. i dq (ma) p1db (dbm) 35 30 25 20 15 10 100 400 200 150 300 350 250 4.5v 4v 3v figure 12. p1db vs. i dq and v ds at 900 mhz. 4.5v 4v 3v i dq (ma) p1db (dbm) 35 30 25 20 15 10 100 400 200 150 300 350 250 figure 13. p1db vs. i dq and v ds at 3.9 ghz. 4.5v 4v 3v i dq (ma) p1db (dbm) 35 30 25 20 15 10 100 400 200 150 300 350 250 figure 14. small signal gain vs i ds and v ds at 2 ghz. i d (ma) gain (dbm) 17 16 15 14 13 12 11 10 100 400 200 150 300 350 250 4.5v 4v 3v figure 15. small signal gain vs i ds and v ds at 900 mhz. i d (ma) gain (dbm) 17 16 15 14 13 12 11 10 100 400 200 150 300 350 250 4.5v 4v 3v figure 16. small signal gain vs i ds and v ds at 3.9 ghz. i d (ma) gain (dbm) 12 11 10 9 8 7 6 5 100 400 200 150 300 350 250 4.5v 4v 3v
6 atf-521p8 typical performance curves , continued (at 25 c unless specified otherwise) tuned for optimal oip3 note: bias current for the above charts are quiescent conditions. actual level may increase depending on amount of rf drive. figure 17. pae @ p1db vs. i dq and v ds at 2 ghz. i dq (ma) pae (%) 70 60 50 40 30 20 10 100 400 200 150 300 350 250 4.5v 4v 3v figure 18. pae @ p1db vs. i dq and v ds at 900 mhz. i dq (ma) pae (%) 70 60 50 40 30 20 10 100 400 200 150 300 350 250 4.5v 4v 3v figure 19. pae @ p1db vs. i dq and v ds at 3.9 ghz. i dq (ma) pae (%) 50 45 40 35 30 25 20 15 10 100 400 200 150 300 350 250 4.5v 4v 3v figure 20. oip3 vs. temp and freq tuned for optimal oip3 at 4.5v, 200 ma. frequency (ghz) oip3 (dbm) 50 45 40 35 30 25 20 15 0.5 4 3.5 1.5 1 2.5 3 2 85 c 25 c -40 c figure 21. p1db vs. temp and freq tuned for optimal oip3 at 4.5v, 200 ma. frequency (ghz) p1db (dbm) 29 27 25 23 21 19 17 15 0.5 4 3.5 1.5 1 2.5 3 2 85 c 25 c -40 c figure 22. gain vs. temp and freq tuned for optimal oip3 at 4.5v, 200 ma. frequency (ghz) gain (db) 20 15 10 5 0 0.5 4 3.5 1.5 1 2.5 3 2 85 c 25 c -40 c figure 23. pae vs temp and freq tuned for optimal oip3 at 4.5v, 200 ma. frequency (ghz) pae (%) 70 60 50 40 30 20 10 0 0.5 4 3.5 1.5 1 2.5 3 2 85 c 25 c -40 c
7 atf-521p8 typical performance curves (at 25 c unless specified otherwise) tuned for optimal p1db note: bias current for the above charts are quiescent conditions. actual level may increase depending on amount of rf drive. figure 24. oip3 vs. i ds and v ds at 2 ghz. i d (ma) oip3 (dbm) 45 40 35 30 25 20 15 10 100 400 200 150 300 350 250 4.5v 4v 3v 200 150 300 350 250 4.5v 4v 3v figure 25. oip3 vs. i ds and v ds at 900 mhz. 4.5v 4v 3v i d (ma) oip3 (dbm) 45 40 35 30 25 20 15 10 100 400 figure 27. p1db vs. i dq and v ds at 2 ghz. 4.5v 4v 3v i dq (ma) p1db (dbm) 35 30 25 20 15 10 100 400 200 150 300 350 250 figure 28. p1db vs. i dq and v ds at 900 mhz. 4.5v 4v 3v i dq (ma) p1db (dbm) 35 30 25 20 15 10 100 400 200 150 300 350 250 figure 29. p1db vs. i dq and v ds at 3.9 ghz. 4.5v 4v 3v i dq (ma) p1db (dbm) 35 30 25 20 15 10 100 400 200 150 300 350 250 figure 30. gain vs i ds and v ds at 2 ghz. 4.5v 4v 3v i d (ma) gain (dbm) 17 15 13 11 9 7 5 100 400 200 150 300 350 250 figure 31. gain vs i ds and v ds at 900 mhz. 4.5v 4v 3v i d (ma) gain (dbm) 17 15 13 11 9 7 5 100 400 200 150 300 350 250 figure 32. gain vs i ds and v ds at 3.9 ghz. i d (ma) gain (dbm) 17 15 13 11 9 7 5 100 400 200 150 300 350 250 4.5v 4v 3v figure 26. oip3 vs. i ds and v ds at 3.9 ghz. 4.5v 4v 3v i d (ma) oip3 (dbm) 50 45 40 35 30 25 20 15 10 100 400 200 150 300 350 250
8 atf-521p8 typical performance curves , continued (at 25 c unless specified otherwise) tuned for optimal p1db note: bias current for the above charts are quiescent conditions. actual level may increase depending on amount of rf drive. figure 33. pae @ p1db vs. i dq and v ds at 2 ghz. i dq (ma) pae (%) 60 55 50 45 40 35 30 25 20 100 400 350 200 150 300 250 4.5v 4v 3v i dq (ma) pae (%) 55 50 45 40 35 30 25 20 100 400 350 200 150 300 250 figure 34. pae @ p1db vs. i dq and v ds at 900 mhz. 4.5v 4v 3v figure 35. pae @ p1db vs. i dq and v ds at 3.9 ghz. i dq (ma) pae (%) 40 35 30 25 20 100 400 200 150 300 350 250 4.5v 4v 3v figure 36. oip3 vs. temp and freq tuned for optimal p1db at 4.5v, 200 ma. frequency (ghz) oip3 (dbm) 50 45 40 35 30 25 20 15 0.5 4 3.5 1.5 1 2.5 3 2 85 c 25 c -40 c figure 38. gain vs. temp and freq tuned for optimal p1db at 4.5v, 200 ma. frequency (ghz) gain (db) 20 15 10 5 0 0.5 4 3.5 1.5 1 2.5 3 2 85 c 25 c -40 c figure 39. pae vs temp and freq tuned for optimal p1db at 4.5v. frequency (ghz) pae (%) 60 50 40 30 20 10 0 0.5 4 85 c 25 c -40 c 3.5 1.5 1 2.5 3 2 figure 37. p1db vs. temp and freq (tuned for optimal p1db at 4.5v, 200 ma). frequency (ghz) p1db (dbm) 32 30 28 26 24 22 20 0.5 4 3.5 1.5 1 2.5 3 2 85 c 25 c -40 c
9 atf-521p8 typical scattering parameters at 25 c, v ds = 4.5v, i ds = 280 ma freq. s 11 s 21 s 12 s 22 msg/mag ghz mag. ang. db mag. ang. db mag. ang. mag. ang. db 0.1 0.613 -96.9 33.2 45.79 141.7 -39.5 0.011 51.3 0.317 -108.3 36.2 0.2 0.780 -131.8 30.0 31.50 121.6 -36.7 0.015 37.1 0.423 -138.5 33.2 0.3 0.831 -147.2 27.3 23.26 111.0 -36.2 0.015 30.6 0.466 -152.4 31.9 0.4 0.855 -156.4 25.1 18.04 104.1 -35.4 0.017 28.2 0.483 -159.9 30.3 0.5 0.860 -162.0 23.5 14.98 99.7 -35.2 0.017 27.4 0.488 -163.8 29.5 0.6 0.878 -166.7 22.0 12.62 95.6 -35.0 0.018 26.1 0.496 -167.0 28.5 0.7 0.888 -170.2 20.8 10.95 92.8 -34.6 0.019 27.4 0.497 -169.9 27.6 0.8 0.887 -172.6 19.7 9.63 90.0 -34.3 0.019 28.9 0.500 -171.7 27.0 0.9 0.894 -174.5 18.7 8.65 87.9 -33.7 0.021 28.5 0.501 -173.6 26.1 1.0 0.886 -177.2 17.9 7.82 85.4 -33.8 0.020 30.3 0.502 -175.7 25.9 1.5 0.892 175.0 14.3 5.20 76.3 -32.8 0.023 34.6 0.502 178.8 23.5 2.0 0.883 168.7 12.1 4.01 68.4 -31.2 0.027 36.7 0.492 173.6 20.2 2.5 0.890 162.8 10.2 3.24 61.5 -30.0 0.032 36.8 0.490 169.8 18.5 3.0 0.884 157.2 8.6 2.71 54.5 -28.9 0.036 39.2 0.494 165.7 16.2 4.0 0.890 146.6 6.1 2.02 40.6 -27.0 0.045 36.1 0.505 157.8 13.8 5.0 0.893 137.0 4.1 1.60 27.6 -25.5 0.053 32.4 0.529 150.3 11.9 6.0 0.896 127.9 2.3 1.31 15.4 -24.2 0.061 28.2 0.551 142.9 10.4 7.0 0.906 119.5 0.9 1.11 3.7 -22.9 0.071 22.9 0.570 135.5 9.6 8.0 0.882 105.6 -0.8 0.92 -9.8 -21.3 0.086 14.5 0.567 127.3 6.8 9.0 0.887 96.4 -1.7 0.82 -22.2 -20.1 0.098 7.2 0.585 117.8 6.2 10.0 0.887 84.6 -2.9 0.72 -33.6 -19.3 0.109 -1.0 0.593 107.3 5.0 11.0 0.882 72.3 -3.9 0.64 -45.8 -18.5 0.119 -10.5 0.617 97.1 3.9 12.0 0.878 62.2 -5.0 0.56 -57.0 -18.0 0.126 -19.8 0.636 86.0 2.8 13.0 0.894 52.0 -6.4 0.48 -67.8 -17.8 0.130 -28.6 0.662 74.7 2.1 14.0 0.888 42.0 -7.6 0.42 -76.2 -17.3 0.137 -36.1 0.697 67.5 0.9 15.0 0.884 34.6 -8.3 0.38 -84.3 -16.6 0.147 -42.9 0.732 58.7 0.3 16.0 0.830 24.7 -9.5 0.34 -92.8 -16.1 0.156 -52.4 0.752 51.9 -1.8 17.0 0.708 11.0 -9.0 0.35 -99.5 -15.4 0.169 -63.8 0.816 46.1 -2.2 18.0 0.790 -12.7 -10.3 0.31 -93.1 -16.4 0.152 -82.8 0.660 41.2 -4.3 freq f min opt opt r n g a ghz db mag. ang. db 0.5 1.20 0.47 170.00 2.8 22.8 1.0 1.30 0.53 -177.00 2.6 20.1 2.0 1.61 0.61 -166.34 2.7 17.3 3.0 1.68 0.69 -155.85 4.0 14.4 4.0 2.12 0.67 -146.98 8.4 11.6 5.0 2.77 0.71 -134.35 19.0 9.9 6.0 2.58 0.79 -125.22 26.7 8.8 7.0 2.85 0.82 -115.35 47.2 7.5 8.0 3.35 0.73 -105.76 65.2 5.7 notes: 1. f min values at 2 ghz and higher are based on measurements while the f mins below 2 ghz have been extrapolated. the f min values are based on a set of 16 noise figure measurements made at 16 different impedances using an atn np5 test system. from these measurements a true f min is calculated. refer to the noise parameter application section for more information. 2. s and noise parameters are measured on a microstrip line made on 0.025 inch thick alumina carrier. the input reference plane is at the end of the gate lead. the output reference plane is at the end of the drain lead. typical noise parameters at 25 c, v ds = 4.5v, i ds = 280 ma figure 40. msg/mag and |s 21 | 2 vs. frequency at 4.5v, 280 ma. mag s 21 frequency (ghz) msg/mag and |s 21 | 2 (db) 020 10 515 40.0 30.0 20.0 10.0 0.0 -10.0 -20.0 msg
10 freq f min opt opt r n g a ghz db mag. ang. db 0.5 0.60 0.30 130.00 2.8 20.2 1.0 0.72 0.35 150.00 2.6 18.4 2.0 0.96 0.47 -175.47 1.9 16.5 3.0 1.11 0.57 -162.03 2.1 13.8 4.0 1.44 0.62 -150.00 4.5 11.2 5.0 1.75 0.69 -136.20 10.0 9.8 6.0 1.99 0.74 -127.35 17.0 8.7 7.0 2.12 0.80 -116.83 28.5 7.5 8.0 2.36 0.69 -108.38 35.6 5.7 notes: 1. f min values at 2 ghz and higher are based on measurements while the f mins below 2 ghz have been extrapolated. the f min values are based on a set of 16 noise figure measurements made at 16 different impedances using an atn np5 test system. from these measurements a true f min is calculated. refer to the noise parameter application section for more information. 2. s and noise parameters are measured on a microstrip line made on 0.025 inch thick alumina carrier. the input reference plane is at the end of the gate lead. the output reference plane is at the end of the drain lead. typical noise parameters, v ds = 4.5v, i ds = 200 ma figure 41. msg/mag and |s 21 | 2 vs. frequency at 4.5v, 200 ma. mag s 21 frequency (ghz) msg/mag and |s 21 | 2 (db) 020 10 515 40.0 30.0 20.0 10.0 0.0 -10.0 -20.0 msg atf-521p8 typical scattering parameters, v ds = 4.5v, i ds = 200 ma freq. s 11 s 21 s 12 s 22 msg/mag ghz mag. ang. db mag. ang. db mag. ang. mag. ang. db 0.1 0.823 -89.9 34.4 52.21 135.6 -37.9 0.013 46.2 0.388 -113.0 36.0 0.2 0.873 -128.7 30.5 33.39 115.7 -35.6 0.017 32.0 0.478 -143.2 32.9 0.3 0.879 -145.5 27.6 23.90 106.3 -34.9 0.018 27.0 0.507 -156.0 31.2 0.4 0.885 -155.1 25.2 18.25 100.5 -34.7 0.018 25.8 0.518 -163.1 30.1 0.5 0.883 -161.1 23.6 15.12 96.6 -34.4 0.019 24.8 0.519 -166.7 29.0 0.6 0.897 -165.9 22.1 12.66 92.9 -34.1 0.020 24.2 0.525 -169.6 28.0 0.7 0.895 -169.5 20.8 10.95 90.5 -33.7 0.021 24.2 0.526 -172.2 27.2 0.8 0.894 -171.9 19.6 9.59 88.0 -33.6 0.021 25.3 0.528 -174.0 26.6 0.9 0.900 -174.7 18.7 8.64 86.2 -33.1 0.022 26.2 0.528 -175.6 25.9 1 0.893 -176.6 17.8 7.78 83.7 -33.1 0.022 27.6 0.529 -177.7 25.5 1.5 0.894 175.3 14.3 5.17 75.7 -32.1 0.025 32.6 0.527 177.2 23.2 2 0.889 168.5 12.0 4.00 67.8 -30.8 0.029 33.6 0.516 172.1 21.4 2.5 0.888 162.6 10.2 3.22 61.3 -29.8 0.032 35.2 0.514 168.1 18.4 3 0.892 157.0 8.6 2.69 54.5 -28.6 0.037 35.6 0.517 164.0 16.7 4 0.884 146.5 6.0 2.00 40.7 -26.8 0.046 34.4 0.526 156.0 13.5 5 0.891 137.0 4.0 1.59 28.3 -25.2 0.055 30.5 0.548 148.3 11.9 6 0.889 127.9 2.3 1.30 16.4 -24.0 0.063 26.4 0.568 141.0 10.1 7 0.902 119.6 0.9 1.11 4.8 -22.8 0.072 21.0 0.584 133.5 9.4 8 0.881 105.6 -0.9 0.90 -8.8 -21.3 0.086 13.3 0.580 124.9 6.7 9 0.891 96.0 -1.7 0.83 -20.1 -20.2 0.098 5.6 0.594 115.8 6.4 10 0.876 83.9 -2.9 0.72 -32.1 -19.3 0.108 -3.2 0.600 105.3 4.6 11 0.885 73.1 -3.6 0.66 -43.7 -18.5 0.119 -12.1 0.622 95.0 4.2 12 0.885 60.9 -4.8 0.57 -54.1 -18.0 0.126 -21.6 0.641 84.1 3.0 13 0.893 53.0 -6.3 0.48 -66.2 -17.7 0.131 -29.9 0.663 73.1 2.1 14 0.889 42.2 -7.2 0.44 -74.0 -17.2 0.138 -36.7 0.698 65.7 1.2 15 0.894 34.3 -7.8 0.41 -80.6 -16.9 0.143 -44.1 0.732 57.4 1.0 16 0.840 25.0 -8.4 0.38 -83.4 -16.2 0.154 -54.3 0.750 51.0 -0.8 17 0.719 9.1 -10.0 0.32 -90.1 -15.4 0.171 -64.8 0.815 44.5 -3.2 18 0.794 -8.1 -12.2 0.25 -102.3 -16.7 0.147 -84.1 0.655 40.4 -5.9
11 freq f min opt opt r n g a ghz db mag. ang. db 0.5 0.60 0.19 162.00 3.0 20.0 1.0 0.72 0.30 164.00 2.6 18.3 2.0 0.81 0.44 176.97 2.0 15.9 3.0 0.92 0.56 -164.98 2.0 13.6 4.0 1.24 0.59 -155.51 3.4 11.1 5.0 1.50 0.70 -136.55 11.1 9.7 6.0 1.60 0.75 -128.59 16.0 8.7 7.0 1.88 0.81 -117.31 24.0 7.6 8.0 2.02 0.68 -109.54 28.8 5.6 typical noise parameters, v ds = 4.5v, i ds = 120 ma figure 42. msg/mag and |s 21 | 2 vs. frequency at 4.5v, 120 ma. mag s 21 frequency (ghz) msg/mag and |s 21 | 2 (db) 020 10 515 40.0 30.0 20.0 10.0 0.0 -10.0 -20.0 msg atf-521p8 typical scattering parameters, v ds = 4.5v, i ds = 120 ma freq. s 11 s 21 s 12 s 22 msg/mag ghz mag. ang. db mag. ang. db mag. ang. mag. ang. db 0.1 0.913 -84.6 34.2 51.26 135.4 -36.4 0.015 49.0 0.423 -106.6 35.3 0.2 0.900 -125.0 30.3 32.80 115.4 -33.9 0.020 31.2 0.499 -139.4 32.1 0.3 0.896 -142.0 27.4 23.39 106.1 -33.4 0.021 25.3 0.522 -153.4 30.5 0.4 0.893 -152.3 25.1 17.89 100.3 -32.9 0.023 23.5 0.530 -161.1 28.9 0.5 0.882 -158.4 23.4 14.75 96.3 -32.6 0.023 22.5 0.531 -165.0 28.1 0.6 0.895 -164.2 21.8 12.36 92.9 -32.7 0.023 20.6 0.537 -168.4 27.3 0.7 0.893 -167.8 20.6 10.71 90.5 -32.4 0.024 20.4 0.537 -171.2 26.5 0.8 0.895 -170.8 19.5 9.39 88.0 -32.3 0.024 21.1 0.539 -173.1 25.9 0.9 0.897 -173.0 18.5 8.44 86.1 -32.2 0.025 22.1 0.539 -174.8 25.3 1 0.895 -175.5 17.6 7.59 83.6 -31.8 0.026 23.0 0.540 -176.9 24.7 1.5 0.893 176.0 14.1 5.07 75.3 -31.1 0.028 25.5 0.538 177.4 22.6 2 0.889 169.2 11.8 3.89 67.8 -30.0 0.032 27.9 0.528 172.2 20.8 2.5 0.882 163.6 10.0 3.15 61.2 -29.0 0.036 30.2 0.526 168.1 19.4 3 0.888 157.9 8.4 2.62 54.6 -28.2 0.039 30.2 0.528 163.9 16.9 4 0.883 146.8 5.9 1.97 40.7 -26.5 0.047 29.7 0.536 155.7 13.6 5 0.885 137.7 3.8 1.55 28.2 -25.2 0.055 26.3 0.556 148.1 11.6 6 0.892 128.0 2.1 1.28 16.7 -24.0 0.063 21.9 0.576 140.5 10.2 7 0.894 120.4 0.6 1.08 5.1 -22.8 0.072 18.2 0.591 133.1 8.9 8 0.880 105.7 -1.0 0.89 -8.7 -21.2 0.087 10.6 0.585 124.3 6.6 9 0.876 96.5 -1.9 0.81 -20.8 -20.1 0.099 3.2 0.602 114.9 5.7 10 0.879 84.4 -3.0 0.71 -32.7 -19.3 0.108 -5.2 0.605 104.5 4.7 11 0.889 72.8 -3.8 0.65 -44.3 -18.6 0.118 -13.5 0.624 94.2 4.3 12 0.881 62.4 -5.2 0.55 -56.0 -18.1 0.125 -23.1 0.642 83.4 2.7 13 0.893 54.0 -6.3 0.48 -66.6 -17.7 0.130 -31.4 0.664 72.4 2.2 14 0.891 42.1 -7.2 0.44 -72.6 -17.3 0.136 -38.4 0.697 65.1 1.2 15 0.888 34.1 -8.3 0.39 -79.2 -16.8 0.144 -45.9 0.732 56.7 0.4 16 0.845 25.3 -9.1 0.35 -89.6 -16.1 0.157 -55.0 0.751 50.4 -1.5 17 0.828 13.2 -11.2 0.28 -95.9 -15.6 0.167 -64.2 0.821 44.0 -3.9 18 0.827 -10.2 -11.0 0.28 -92.5 -16.6 0.147 -86.1 0.654 39.9 -4.3 notes: 1. f min values at 2 ghz and higher are based on measurements while the f mins below 2 ghz have been extrapolated. the f min values are based on a set of 16 noise figure measurements made at 16 different impedances using an atn np5 test system. from these measurements a true f min is calculated. refer to the noise parameter application section for more information. 2. s and noise parameters are measured on a microstrip line made on 0.025 inch thick alumina carrier. the input reference plane is at the end of the gate lead. the output reference plane is at the end of the drain lead.
12 freq f min opt opt r n g a ghz db mag. ang. db 0.5 0.67 0.21 155.00 2.8 20.1 1.0 0.74 0.30 164.00 2.6 18.4 2.0 0.96 0.46 -176.61 2.1 16.4 3.0 1.24 0.57 -162.19 2.8 13.9 4.0 1.44 0.62 -152.18 4.5 11.4 5.0 1.62 0.69 -135.43 10.0 10.0 6.0 1.83 0.74 -127.94 17.0 8.7 7.0 1.99 0.82 -117.20 27.7 7.7 8.0 2.21 0.71 -108.96 35.3 5.9 typical noise parameters, v ds = 4v, i ds = 200 ma figure 43. msg/mag and |s 21 | 2 vs. frequency at 4v, 200 ma. mag s 21 frequency (ghz) msg/mag and |s 21 | 2 (db) 020 10 515 40.0 30.0 20.0 10.0 0.0 -10.0 -20.0 msg atf-521p8 typical scattering parameters, v ds = 4v, i ds = 200 ma freq. s 11 s 21 s 12 s 22 msg/mag ghz mag. ang. db mag. ang. db mag. ang. mag. ang. db 0.1 0.843 -90.5 34.3 51.89 134.8 -37.7 0.013 46.5 0.408 -118.1 36.0 0.2 0.879 -129.3 30.3 32.88 115.0 -35.4 0.017 32.1 0.507 -146.1 32.9 0.3 0.888 -146.1 27.4 23.48 105.8 -35.1 0.018 26.0 0.539 -158.3 31.2 0.4 0.892 -155.6 25.1 17.91 100.1 -34.4 0.019 25.1 0.549 -164.8 29.7 0.5 0.886 -161.5 23.4 14.80 96.3 -34.2 0.020 24.6 0.551 -168.2 28.7 0.6 0.896 -165.7 21.8 12.37 92.7 -34.2 0.020 24.1 0.556 -170.9 27.9 0.7 0.897 -169.5 20.6 10.74 90.5 -33.6 0.021 24.7 0.557 -173.5 27.1 0.8 0.898 -172.2 19.5 9.39 88.1 -33.5 0.021 24.4 0.559 -175.2 26.5 0.9 0.896 -174.9 18.6 8.47 85.9 -33.3 0.022 26.5 0.559 -176.9 25.9 1 0.896 -176.7 17.6 7.61 84.0 -32.9 0.023 26.3 0.560 -178.7 25.2 1.5 0.898 175.2 14.1 5.06 75.7 -32.1 0.025 29.9 0.558 176.0 23.1 2 0.887 168.0 11.8 3.91 68.1 -30.7 0.029 35.2 0.547 170.9 21.3 2.5 0.893 162.8 10.0 3.15 61.7 -29.5 0.034 35.8 0.545 166.9 18.9 3 0.886 156.9 8.4 2.63 55.1 -28.4 0.038 35.8 0.547 162.6 16.3 4 0.887 146.6 5.9 1.97 41.5 -26.7 0.046 33.2 0.554 154.3 13.6 5 0.894 136.8 3.9 1.57 29.4 -25.1 0.056 29.6 0.572 146.6 11.9 6 0.898 127.4 2.1 1.28 17.7 -23.9 0.064 25.5 0.590 139.0 10.3 7 0.896 119.7 0.7 1.09 6.3 -22.6 0.074 20.4 0.603 131.6 8.9 8 0.879 105.4 -0.9 0.90 -7.1 -21.1 0.088 12.4 0.594 122.7 6.6 9 0.888 95.0 -1.7 0.82 -19.3 -20.1 0.099 4.7 0.609 113.2 6.1 10 0.872 84.1 -2.9 0.72 -30.9 -19.2 0.110 -4.3 0.610 102.9 4.4 11 0.880 72.4 -3.8 0.65 -42.8 -18.6 0.118 -12.9 0.629 92.6 3.8 12 0.875 60.4 -4.8 0.58 -53.3 -18.0 0.126 -22.8 0.647 81.9 2.8 13 0.908 52.4 -6.2 0.49 -63.4 -17.7 0.130 -31.4 0.666 71.0 2.6 14 0.898 41.3 -7.1 0.44 -73.5 -17.2 0.138 -38.0 0.699 64.0 1.5 15 0.888 34.1 -8.2 0.39 -80.2 -16.8 0.144 -45.6 0.734 55.9 0.5 16 0.815 24.1 -8.9 0.36 -85.3 -16.2 0.156 -54.7 0.750 49.3 -1.7 17 0.725 11.3 -9.9 0.32 -90.9 -15.5 0.167 -66.0 0.809 43.5 -3.1 18 0.792 -9.8 -10.2 0.31 -95.1 -16.6 0.147 -84.8 0.652 39.7 -4.2 notes: 1. f min values at 2 ghz and higher are based on measurements while the f mins below 2 ghz have been extrapolated. the f min values are based on a set of 16 noise figure measurements made at 16 different impedances using an atn np5 test system. from these measurements a true f min is calculated. refer to the noise parameter application section for more information. 2. s and noise parameters are measured on a microstrip line made on 0.025 inch thick alumina carrier. the input reference plane is at the end of the gate lead. the output reference plane is at the end of the drain lead.
13 atf-521p8 typical scattering parameters, v ds = 3v, i ds = 200 ma freq. s 11 s 21 s 12 s 22 msg/mag ghz mag. ang. db mag. ang. db mag. ang. mag. ang. db 0.1 0.867 -94.6 33.7 48.20 132.4 -36.8 0.014 45.1 0.482 -132.4 35.4 0.2 0.894 -132.9 29.4 29.66 113.2 -34.9 0.018 28.5 0.601 -154.2 32.2 0.3 0.899 -148.2 26.5 21.06 104.4 -34.1 0.020 23.2 0.636 -163.8 30.2 0.4 0.896 -157.2 24.1 16.00 99.1 -34.0 0.020 23.7 0.647 -169.2 29.0 0.5 0.892 -162.8 22.4 13.20 95.6 -33.6 0.021 24.5 0.650 -171.9 28.0 0.6 0.910 -167.4 20.8 11.00 92.3 -33.2 0.022 22.9 0.655 -174.4 27.0 0.7 0.906 -170.8 19.6 9.51 90.2 -33.2 0.022 23.9 0.657 -176.7 26.4 0.8 0.902 -173.6 18.4 8.35 87.8 -33.0 0.022 24.6 0.658 -178.2 25.8 0.9 0.907 -175.2 17.5 7.51 86.3 -32.9 0.023 27.0 0.660 -179.5 25.1 1 0.902 -177.7 16.6 6.76 84.2 -32.5 0.024 26.9 0.659 178.6 24.5 1.5 0.900 174.2 13.1 4.50 76.4 -31.5 0.027 32.7 0.656 173.4 22.2 2 0.896 168.1 10.8 3.49 69.1 -29.9 0.032 32.9 0.647 167.9 20.4 2.5 0.896 162.3 9.0 2.82 63.0 -29.0 0.036 34.3 0.642 163.7 18.6 3 0.887 156.7 7.4 2.35 56.9 -27.7 0.041 35.0 0.643 159.2 15.6 4 0.890 145.7 4.9 1.76 43.8 -26.1 0.050 32.2 0.645 150.4 12.9 5 0.898 136.3 3.0 1.41 32.1 -24.5 0.059 28.3 0.659 142.1 11.3 6 0.896 127.4 1.3 1.16 21.6 -23.4 0.068 23.5 0.671 134.3 9.5 7 0.904 119.4 -0.2 0.98 10.3 -22.1 0.078 17.7 0.677 126.6 8.5 8 0.877 104.9 -1.6 0.83 -2.3 -20.7 0.092 9.0 0.651 117.0 5.9 9 0.883 94.8 -2.4 0.76 -13.0 -19.8 0.102 1.3 0.661 107.2 5.3 10 0.877 83.1 -3.5 0.67 -26.0 -18.9 0.113 -7.3 0.657 96.8 4.0 11 0.875 71.7 -4.4 0.60 -36.3 -18.3 0.121 -16.6 0.670 86.7 3.1 12 0.863 60.6 -5.4 0.54 -47.4 -17.8 0.128 -25.1 0.680 76.2 1.9 13 0.910 51.6 -6.5 0.47 -57.9 -17.6 0.132 -33.6 0.694 65.9 2.3 14 0.868 40.9 -7.5 0.42 -62.8 -17.2 0.138 -40.4 0.721 59.3 0.2 15 0.863 33.4 -8.1 0.39 -74.7 -16.8 0.144 -47.6 0.748 51.3 -0.2 16 0.835 25.2 -9.6 0.33 -78.2 -16.3 0.154 -56.8 0.758 44.9 -2.1 17 0.720 11.2 -9.5 0.33 -90.8 -15.8 0.161 -67.6 0.818 39.4 -2.6 18 0.780 -7.7 -11.6 0.26 -92.8 -17.0 0.142 -85.1 0.655 37.1 -5.7 freq f min opt opt r n g a ghz db mag. ang. db 0.5 0.66 0.22 147.00 2.9 20.0 1.0 0.72 0.30 160.00 2.6 18.3 2.0 0.87 0.42 -179.94 1.9 16.0 3.0 1.00 0.59 -163.63 1.6 13.7 4.0 1.32 0.63 -153.81 3.7 11.3 5.0 1.49 0.72 -135.10 10.0 9.9 6.0 1.59 0.74 -128.97 15.0 8.5 7.0 1.79 0.78 -117.68 25.1 7.6 8.0 1.96 0.70 -110.04 29.2 5.6 typical noise parameters, v ds = 3v, i ds = 200 ma figure 44. msg/mag and |s 21 | 2 vs. frequency at 3v, 200 ma. mag s 21 frequency (ghz) msg/mag and |s 21 | 2 (db) 020 10 515 40.0 30.0 20.0 10.0 0.0 -10.0 -20.0 msg notes: 1. f min values at 2 ghz and higher are based on measurements while the f mins below 2 ghz have been extrapolated. the f min values are based on a set of 16 noise figure measurements made at 16 different impedances using an atn np5 test system. from these measurements a true f min is calculated. refer to the noise parameter application section for more information. 2. s and noise parameters are measured on a microstrip line made on 0.025 inch thick alumina carrier. the input reference plane is at the end of the gate lead. the output reference plane is at the end of the drain lead.
14 atf-521p8 applications information description agilent s atf-521p8 is an enhancement mode phemt designed for high linearity and medium power applications. with an oip3 of 42 dbm and a 1db compression point of 26 dbm, atf-521p8 is well suited as a base station transmit driver or a first or second stage lna in a receive chain. whether the design is for a w-cdma, cdma, or gsm basestation, this device delivers good linearity in the form of oip3 or aclr, which is required for standards with high peak to average ratios. application guidelines the atf-521p8 device operates as a normal fet requiring input and output matching as well as dc biasing. unlike a depletion mode transistor, this enhance- ment mode device only requires a single positive power supply, which means a positive voltage is placed on the drain and gate in order for the transistor to turn on. this application note walks through the rf and dc design employed in a single fet ampli- fier. included in this description is an active feedback scheme to accomplish this dc biasing. rf input & output matching in order to achieve maximum linearity, the appropriate input ( s ) and output ( l ) impedances must be presented to the device. correctly matching from these impedances to 50 ? s will result in maximum linearity. although atf-521p8 may be used in other impedance systems, data col- lected for this data sheet is all referenced to a 50 ? system. the input load pull parameter at 2 ghz is shown in figure 1 along with the optimum s11 conjugate match. 16 db 5 db 9 db 3 db s11 * return loss s figure 1. input match for atf-521p8 at 2 ghz. thus, it should be obvious from the illustration above that if this device is matched for maximum return loss i.e. s11*, then oip3 will be sacrificed. conversely, if atf-521p8 is matched for maximum linearity, then return loss will not be greater than 10 db. for most applications, a designer requires vswr greater than 2:1, hence limiting the input match close to s11*. normally, the input return loss of a single ended amplifier is not critical as most basestation lna and driver amplifiers are in a balanced configuration with 90 (quadra- ture) couplers. proceeding from the same premise, the output match of this device becomes much simpler. as background information, it is important to note that oip3 is largely dependant on the output match and that output return loss is also required to be greater than 10 db. so, figure 2 shows how both good output return loss and good linearity could be achieved simultaneously with the same impedance point. of course, these points are valid only at 2 ghz, and other frequen- cies will follow the same design rules but will have different locations. also, the location of these points is largely due to the manufacturing process and partly due to ic layout, but in either case beyond the scope of this application note. s22* l figure 2. output match at 2 ghz. once a designer has chosen the proper input and output imped- ance points, the next step is to choose the correct topology to accomplish this match. for example to perform the above output impedance transforma- tion from 50 ? to the given load parameter of 0.53 -176 , two possible solutions exist. the first potential match is a high pass configuration accomplished by a shunt inductor and a series capacitor shown in figure 3 along with its frequency response in figure 4. c1 rf in rf out l1 figure 3. high pass circuit topology. amp frequency figure 4. high pass frequency response.
15 the second solution is a low pass configuration with a shunt capacitor and a series inductor shown in figure 5 and 6. l1 rf in rf ou t c1 figure 5. low pass circuit topology. amp frequency figure 6. low pass frequency response. the actual values of these components may be calculated by hand on a smith chart or more accurately done on simulation software such as ads. there are some advantages and disadvan- tages of choosing a high pass versus a low pass. for instance, a high pass circuit cuts off low frequency gain, which narrows the usable bandwidth of the amplifier, but consequently helps avoid potential low frequency instability problems. a low pass match offers a much broader frequency response, but it has two major disadvantages. first it has the potential for low fre- quency instability, and second it creates the need for an extra dc blocking capacitor on the input in order to isolate the device gate from the preceding stages. figure 7 displays the input and output matching selected for atf-521p8. in this example the input and output match both essentially function as high pass filters, but the high frequency gain of the device rolls off precipitously giving a narrow band frequency response, yet still wide enough to accommodate a cdma or wcdma transmit band. for more information on rf matching techniques refer to mga-53543 application note. passive bias [1] once the rf matching has been established, the next step is to dc bias the device. a passive biasing example is shown in figure 8. in this example the voltage drop across resistor r3 sets the drain current (id) and is calculated by the following equation: r3 = v dd C v ds (1) p i ds + i bb where, v dd is the power supply voltage; v ds is the device drain to source voltage; i ds is the device drain to source current; i bb for dc stability is 10x the typical gate current; a voltage divider network with r1 and r2 establishes the typical gate bias voltage (vg). r1 = v g (2) p i bb r2 = (v dd C v g ) x r1 (3) p v g often the series resistor, r4, is added to enhance the low fre- quency stability. the complete passive bias example may be found in reference [1]. input output zo c1 c4 zo c5 c6 vdd r3 l4 l1 r4 r5 c3 c2 r1 r2 q1 i b figure 8. passive biasing. + += input match amp frequency amp atf-521p8 frequency frequency amp output match total response amp frequency zo zo 52 c2 c1 c3 l1 rf ou t rf in figure 7. input and output match for atf-521p8 at 2 ghz.
16 active bias [2] due to very high dc power dissipation and small package constraints, it is recommended that atf-521p8 use active biasing. the main advantage of an active biasing scheme is the ability to hold the drain to source current constant over a wide range of temperature variations. a very inexpensive method of accomplishing this is to use two pnp bipolar transistors arranged in a current mirror configuration as shown in figure 9. due to resistors r1 and r3, this circuit is not acting as a true current mirror, but if the voltage drop across r1 and r3 is kept identi- cal then it still displays some of the more useful characteristics of a current mirror. for example, transistor q1 is configured with its base and collector tied together. this acts as a simple pn junction, which helps tempera- ture compensate the emitter- base junction of q2. to calculate the values of r1, r2, r3, and r4 the following param- eters must be know or chosen first: i ds is the device drain-to-source current; i r is the reference current for active bias; v dd is the power supply voltage available; v ds is the device drain-to-source voltage; v g is the typical gate bias; v be1 is the typical base-emitter turn on voltage for q1 & q2; therefore, resistor r3, which sets the desired device drain current, is calculated as follows: r3 = v dd C v ds (4) p i ds + i c2 where, i c2 is chosen for stability to be 10 times the typical gate current and also equal to the reference current i r . the next three equations are used to calculate the rest of the biasing resistors for figure 9. note that the voltage drop across r1 must be set equal to the voltage drop across r3, but with a current of i r . r1 = v dd C v ds (5) p i r r2 sets the bias current through q1. r2 = v ds C v be1 (6) p i r r4 sets the gate voltage for atf-521p8. r4 = v g (7) p i c2 thus, by forcing the emitter voltage (v e ) of transistor q1 equal to v ds , this circuit regulates the drain current similar to a current mirror. as long as q2 operates in the forward active mode, this holds true. in other words, the collector-base junc- tion of q2 must be kept reversed biased. c1 rf in rf ou t l4 l1 l2 l3 r6 r5 r3 r4 c4 c3 c7 c8 c6 c5 q2 c2 r1 r2 q1 v e v g v ds v dd 2 7 atf-521p8 2pl figure 9. active bias circuit.
17 pcb layout a recommended pcb pad layout for the leadless plastic chip carrier (lpcc) package used by the atf-521p8 is shown in figure 10. this layout provides plenty of plated through hole vias for good thermal and rf ground- ing. it also provides a good transition from microstrip to the device package. for more de- tailed dimensions refer to section 9 of the data sheet. figure 10. microstripline layout. rf grounding unlike sot packages, atf-521p8 is housed in a leadless package with the die mounted directly to the lead frame or the belly of the package shown in figure 11. pin 8 drain pin 6 pin 5 source gate pin 3 source bottom view figure 11. lpcc package for atf-521p8. this simplifies rf grounding by reducing the amount of induc- tance from the source to ground. it is also recommended to ground pins 1 and 4 since they are also connected to the device source. pins 3, 5, 6, and 8 are not con- nected, but may be used to help dissipate heat from the package or for better alignment when soldering the device. this three-layer board (figure 12) contains a 10-mil layer and a 52-mil layer separated by a ground plane. the first layer is getek rg200d material with dielectric constant of 3.8. the second layer is for mechanical rigidity and consists of fr4 with dielectric constant of 4.2. high linearity tx driver the need for higher data rates and increased voice capacity gave rise to a new third generation standard know as wideband cdma or umts. this new standard requires higher perfor- mance from radio components such as higher dynamic range and better linearity. for example, a wcdma waveform has a very high peak to average ratio which forces amplifiers in a transmit chain to have very good adjacent channel leakage power ratio or aclr, or else operate in a backed off mode. if the amplifier is not backed off then the wave- form is compressed and the signal becomes very nonlinear. this application example pre- sents a highly linear transmit drive for use in the 2.14ghz frequency range. using the rf matching techniques described earlier, atf-521p8 is matched to the following input and output impedances: input match output match 2pl 50 ohm s11* = 0.89 -169 l = 0.53 -176 50 ohm figure 13. atf-521p8 matching. figure 12. atf-521p8 demoboard. p j1 j2 bcv62b c1 c8 0 0 l3 r1 r2 r3 r4 r5 r6 l4 c2 c3 c5 c6 c4 c7 l2 l1 short
18 as described previously the input impedance must be matched to s11* in order to guarantee return loss greater than 10 db. a high pass network is chosen for this match. the output is matched to l with another high pass network. the next step is to choose the proper dc biasing conditions. from the data sheet, atf-521p8 produces good linearity at a drain current of 200ma and a drain to source voltage of 4.5v. thus to construct the active bias circuit described, the following parameters are given: ids = 200 ma i r = 10 ma v dd = 5 v v ds = 4.5 v v g = 0.62 v v be1 = 0.65 v using equations 4, 5, 6, and 7, the biasing resistor values are calculated in column 2 of table 1, and the actual values used are listed in column 3. resistor calculated actual r1 50 ? 49.9 ? r2 385 ? 383 ? r3 2.38 ? 2.37 ? r4 62 ? 61.9 ? table 1. resistors for active bias. the entire circuit schematic for a 2.14 ghz tx driver amplifier is shown below in figure 14. capacitors c4, c5, and c6 are added as a low frequency bypass. these terminate second order harmonics and help improve linearity. resistors r5 and r6 also help terminate low frequencies, and can prevent resonant frequencies between the two bypass capacitors. performance of atf-521p8 at 2140 mhz atf-521p8 delivers excellent performance in the wcdma frequency band. with a drain-to- source voltage of 4.5v and a drain current of 200 ma, this device has 16.5 db of gain and 1.55 db of noise figure as show in figure 15. frequency (ghz) gain and nf (db) 1.6 20 15 10 5 0 2.6 1.8 2.2 2.4 2.0 gain nf figure 15. gain and noise figure vs. frequency. input and output return loss are both greater that 10 db. although somewhat narrowband, the response is adequate in the frequency range of 2110 mhz to 2170 mhz for the wcdma downlink. if wider band response is need, using a balanced configu- ration improves return loss and doubles oip3. frequency (ghz) input and output return loss (db) 1.6 0 -5 -10 -15 2.6 1.8 2.2 2.4 2.0 s11 s22 figure 16. input and output return loss vs. frequency. perhaps the most critical system level specification for the atf-521p8 lies in its distortion- less output power. typically, amplifiers are characterized for linearity by measuring oip3. this is a two-tone harmonic measure- ment using cw signals. but because wcdma is a modulated waveform spread across 3.84 mhz, it is difficult to corre- lated good oip3 to good aclr. thus, both are measured and presented to avoid ambiguity. figure 14. 2140 mhz schematic. c1=1.2pf rf in rf ou t l4=3.9nh l1=1.0nh l2=12nh l3=39nh r6=1.2 ? r5=10 ? r3=2.37 ? r4=61.9 ? c4=1 f c3=4.7pf c8=1.5pf c7=150pf c5=1 f c6=.1 f q2 i c2 i r c2=1.5nh r1=49.9 ? r2=383 ? q1 v be1 + v g v ds +5v 2 7 atf-521p8 2pl
19 frequency (mhz) oip3 (dbm) 2060 45 40 35 30 25 2200 2080 2120 2140 2100 2160 2180 figure 17. oip3 vs. frequency in wcdma band (pout = 12 dbm). pout (dbm) aclr (db) -3 -30 -35 -40 -45 -50 -55 -60 -65 22 21217 7 figure 18. aclr vs. pout at 5 mhz offset. c1=1.2 pf phycomp 0402cg129c9b200 c2,c8=1.5 pf phycomp 0402cg159c9b200 c3=4.7 pf phycomp 0402cg479c9b200 c4,c6=.1 f phycomp 06032f104m8b200 c5=1 f avx 0805zc105katza c7=150 pf phycomp 0402cg151j9b200 l1=1.0 nh toko ll1005-fh1n0s l2=12 nh toko ll1005-fs12n l3=39 nh toko ll1005-fs39 l4=3.9 nh toko ll1005-fh3n9s r1=49.9 ? rohmrk73h1j49r9f r2=383 ? rohm rk73h1j3830f r3=2.37 ? rohm rk73h1j2r37f r4=61.9 ? rohm rk73h1j61r9f r5=10 ? rohm rk73h1j10r0f r6=1.2 ? rohm rk73h1j1r21f q1, q2 philips bcv62c j1, j2 142-0701-851 table 2. 2140 mhz bill of material. using the 3gpp standards document release 1999 version 2002-6, the following channel configuration was used to test aclr. this table contains the power levels of the main chan- nels used for test model 1. note that the dpch can be made up of 16, 32, or 64 separate channels each at different power levels and timing offsets. for a listing of power levels, channelization codes and timing offset see the entire 3gpp ts 25.141 v3.10.0 (2002-06) standards document at: http://www.3gpp.org/specs/ specs.htm 3gpp ts 25.141 v3.10.0 (2002-06) type pwr (db) p-ccpch+sch -10 primary cpich -10 pich -18 s-ccpch containing pch -18 (sf=256) dpch-64ch -1.1 (sf=128) table 3. aclr channel power configuration. thermal design when working with medium to high power fet devices, thermal dissipation should be a large part of the design. this is done to ensure that for a given ambient temperature the transistor s channel does not exceed the maximum rating, t ch , on the data sheet. for example, atf-521p8 has a maximum channel temperature of 150 c and a channel to board thermal resistance of 45 c/w, thus the entire thermal design hinges from these key data points. the question that must be answered is whether this device can operate in a typical environment with ambient temperature fluctuations from -25 c to 85 c. from figure 19, a very useful equation is derived to calculate the temperature of the channel for a given ambient temperature. these calculations are all incor- porated into agilent technolo- gies appcad. ch-b tch (channel) tb (board or belly of the part) ta ( ambient ) ts (sink) pdiss = vds x ids b-s s-a figure 19. equivalent circuit for thermal resistance. hence very similar to ohms law, the temperature of the channel is calculated with equation 8 below. t ch = p diss ( ch C b + b C s + s C a ) + t amb (8) if no heat sink is used or heat sinking is incorporated into the pcb board then equation 8 may be reduced to: t ch = p diss ( ch C b + b C a ) + t amb (9) where, b C a is the board to ambient thermal resistance; ch C b is the channel to board thermal resistance. the board to ambient thermal resistance thus becomes very important for this is the designer s major source of heat control. to demonstrate the influence of b-a, thermal resis- tance is measured for two very different scenarios using the atf-521p8 demoboard. the first case is done with just the demoboard by itself. the second case is the atf demoboard
20 mounted on a chassis or metal casing, and the results are given below: atf demoboard b-a pcb 1/8" chassis 10.4 c/w pcb no heatsink 32.9 c/w table 4. thermal resistance measurements. therefore calculating the tem- perature of the channel for these two scenarios gives a good indication of what type of heat sinking is needed. case 1: chassis mounted @ 85 c tch = p x ( ch-b + b-a ) + ta =.9w x (45+10.4) c/w +85 c tch = 135 c case 2: no heatsink @ 85 c tch = p x ( ch-b + b-a ) + ta =.9w x (45+32.9) c/w + 85 c tch = 155 c in other words, if the board is mounted to a chassis, the chan- nel temperature is guaranteed to be 135 c safely below the 150 c maximum. but on the other hand, if no heat sinking is used and the b-a is above 27 c/w (32.9 c/w in this case), then the power must be derated enough to lower the temperature below 150 c. this can be better under- stood with figure 20 below. note power is derated at 13 mw/ c for the board with no heat sink and no derating is required for the chassis mounted board until an ambient temperature of 100 c. pdiss (w) 0.9w 0 81 100 150 tamb ( c ) no heatsink (13 mw/ c) mounted on chassis (18 mw/ c) figure 20. derating for atf- 521p8. thus, for reliable operation of atf-521p8 and extended mtbf, it is recommended to use some form of thermal heatsinking. this may include any or all of the following suggestions: ? maximize vias underneath and around package; ? maximize exposed surface metal; ? use 1 oz or greater copper clad; ? minimize board thickness; ? metal heat sinks or extrusions; ? fans or forced air; ? mount pcb to chassis. summary a high linearity tx driver amplifier for wcdma has been presented and designed using agilent s atf-521p8. this includes rf, dc and good ther- mal dissipation practices for reliable lifetime operation. a summary of the typical perfor- mance for atf-521p8 demoboard at 2140 mhz is as follows: demo board results at 2140 mhz gain 16.5 db oip3 41.2 dbm aclr -58 dbc p1db 24.8 dbm nf 1.55 db references [1] ward, a. (2001) agilent atf-54143 low noise enhance- ment mode pseudomorphic hemt in a surface mount plastic package, 2001 [internet], available from: [accessed 22 august, 2002]. [2] biasing circuits and considerations for gaas mesfet power amplifiers, 2001 [internet], available from: [accessed 22 august, 2002]
21 2 x 2 lpcc (jedec dfp-n) package dimensions ordering information part number no. of devices container atf-521p8-tr1 3000 7 reel atf-521p8-tr2 10000 13 reel ATF-521P8-BLK 100 antistatic bag device models refer to agilent s web site www.agilent.com/view/rf d e 8 7 6 5 a d1 e1 p e pin1 r l b dimensions are in millimeters dimensions min. 0.70 0 0.225 1.9 0.65 1.9 1.45 nom. 0.75 0.02 0.203 ref 0.25 2.0 0.80 2.0 1.6 0.50 bsc max. 0.80 0.05 0.275 2.1 0.95 2.1 1.75 symbol a a1 a2 b d d1 e e1 e 1 pin1 2 3 4 2px top view end view end view bottom view a2 a a1
22 device orientation pcb land pattern and stencil design 2.80 (110.24) 0.70 (27.56) 0.25 (9.84) 0.25 (9.84) 0.50 (19.68) 0.28 (10.83) 0.60 (23.62) 0.20 (7.87) pin 1 solder mask rf transmission line 0.80 (31.50) 0.15 (5.91) 0.55 (21.65) 1.60 (62.99) + 2.72 (107.09) 0.63 (24.80) 0.22 (8.86) 0.32 (12.79) 0.50 (19.68) 0.25 (9.74) 0.63 (24.80) stencil layout (top view) pcb land pattern (top view) 0.72 (28.35) pin 1 1.54 (60.61) user feed direction cover tape carrier tape reel 8 mm 4 mm 2px 2px 2px 2px
23 tape dimensions p 0 p f w d 1 e p 2 a 0 10 max t 1 k 0 description symbol size (mm) size (inches) length width depth pitch bottom hole diameter a 0 b 0 k 0 p d 1 2.30 0.05 2.30 0.05 1.00 0.05 4.00 0.10 1.00 + 0.25 0.091 0.004 0.091 0.004 0.039 0.002 0.157 0.004 0.039 + 0.002 cavity diameter pitch position d p 0 e 1.50 0.10 4.00 0.10 1.75 0.10 0.060 0.004 0.157 0.004 0.069 0.004 perforation width thickness w t 1 8.00 + 0.30 0.254 0.02 0.315 0.012 8.00 C 0.10 0.315 0.004 0.010 0.0008 carrier tape cavity to perforation (width direction) cavity to perforation (length direction) f p 2 3.50 0.05 2.00 0.05 0.138 0.002 0.079 0.002 distance width tape thickness c t t 5.4 0.10 0.062 0.001 0.205 0.004 0.0025 0.0004 cover tape d ++ t t 10 max b 0
www.agilent.com/semiconductors for product information and a complete list of distributors, please go to our web site. for technical assistance call: americas/canada: +1 (800) 235-0312 or (916) 788 6763 europe: +49 (0) 6441 92460 china: 10800 650 0017 hong kong: (+65) 6271 2451 india, australia, new zealand: (+65) 6271 2394 japan: (+81 3) 3335-8152(domestic/international), or 0120-61-1280(domestic only) korea: (+65) 6271 2194 malaysia, singapore: (+65) 6271 2054 taiwan: (+65) 6271 2654 data subject to change. copyright ? 2003 agilent technologies, inc. obsoletes 5988-8403 july 29, 2003 5988-9974en

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